Digital implementation of spread spectrum communications system

ABSTRACT

The present invention comprises the digital implementation of a spread spectrum communications system including a spread spectrum modulator or transmitter and a spread spectrum receiver or demodulator. The demodulator, as modified with optional techniques, operates at zero I. Under these circumstances the receiver is susceptible to 1/F noise, as well as transient overloads. The addition of a high pass filter stage to only the receiver, in a communication system, without modifying the transmitter, will be recognized by those skilled in the art as violating the basic precepts of communications which attempt to tailor the characteristics of a receiver and transmitter to each other. However the applicant has found that demodulating RZ encoded data with an added high pass filter results in the production of Manchester data which is one of the forms of data encoding that can be handled by the demodulator.

This is a divisional of application Ser. No. 08/081,690 filed on Jun.25, 1993, now U.S. Pat. No. 5,331,446.

TECHNICAL FIELD

The present invention relates to spread spectrum communications systems.

1. Related Applications

Co-pending application Ser. No. 08/081,689, filed Jun. 25, 1993discloses synchronization of transmit and receive events in ahalf-duplex "ping-pong" radio link.

Co-pending application Ser. No. 08/084,978, filed Jun. 25, 1993discloses a minimal logic correlator.

Co-pending application Ser. No. 08/081,939, filed Jun. 25, 1993discloses a method and apparatus for synchronization between real-timesampled audio applications operating full-duplex over a half-duplexradio link.

Each of the above-identified applications is assigned to the assignee ofthis application and the subject matter incorporated herein by thisreference.

2. Background

An important criterion of spread spectrum operation is resistance tomulti-path interference. It is well known (see Dixon, Spread SpectrumSystems, 2nd Edition, page 275, 1984) that direct sequence spreadsystems exhibit resistance to multi-path for differential delays greaterthan a chip width. This resistance results from the fact that a signalwhich is delayed by more than a chip width is no longer correlated witha direct or desired signal and, presumably, is rejected as interferenceby a correlator which is aligned to the timing of the direct or desiredsignal.

In urban and indoor radio systems, multi-path differential delay spreadsrange from 10 to 125 nanoseconds in homes and offices. In large,enclosed arenas, exhibit halls and sports pavilions, delay spreads rangefrom 25 nanoseconds to over one μsec. Outdoors in urban areas, delayspreads can range from 50 nanoseconds to over three μsec. In order for aconventional direct sequence spread spectrum communications system toexhibit significant multi-path resistance over the entire range listedabove (i.e. differential delays spread less than a chip width), it mustoperate at chip rates on the order of 50-100 megachips per second. Incommercial applications, this is not practical because of bandwidthrestrictions and digital logic speed limitations. The present inventionprovides a different approach, namely, employing modulation anddemodulation schemes that inherently have resistance to multi-path andwhose performance is enhanced by incorporation of direct sequence spreadspectrum techniques. In this regard, see McIntosh U.S. Pat. No.4,862,478 which describes systems exhibiting inherent resistance tomulti-path interference. The present invention is directed at achievingthe same end, i.e. resistance to multi-path interference. In contrast tothe system described in U.S. Pat. No. 4,862,478, the present inventionimproves on the manufactureability. In particular, the present inventionimproves on the system of U.S. Pat. No. 4,862,478 in two areas. As isdescribed in detail hereinafter, the demodulator or receiver operates atzero IF. In addition, by using an A/D converter prior to recovering thebaseband modulation, the delays can be digitally implemented.

SUMMARY OF THE INVENTION

The present invention comprises a digital implementation of a spreadspectrum communications system including a spread spectrum modulator ortransmitter and a spread spectrum receiver or demodulator. FIG. 1 showsa demodulator which can be used with the modulator of FIGS. 2, 3 or 4.FIGS. 5 and 7 show optional variations on the demodulator of FIG. 1.

As is described below, the demodulator of FIG. 1 or that demodulator asmodified with the optional techniques of FIGS. 5 and 7, operates at zeroIF. Under these circumstances, the receiver is susceptible to 1/F noise,as well as transient overloads. Adding a high pass filter in I and Qchannels, as shown in FIG. 6, is advantageous. As is describedhereinafter, the addition of a high pass filter stage to only thereceiver in a communications system, without modifying the transmitter,will be recognized by those skilled in the art as violating basicprecepts of communications which attempt to tailor the characteristicsof a receiver and transmitter to each other. However, applicant hasfound that demodulating RZ encoded data with an added high pass filteras shown in FIG. 6, results in the production of Manchester encoded datawhich is one of the forms of data encoding that can be handled by thedemodulator.

BRIEF DESCRIPTION OF THE DRAWINGS

The present invention will now be described in the following portions ofthis specification when taken in conjunction with the attached drawingsin which:

FIG. 1 is a block diagram of a suitable demodulator;

FIGS. 2, 3 and 4 illustrate suitable forms of a modulator;

FIG. 5 illustrates an alternative demodulator structure, alternative tothat of FIG. 1;

FIG. 6 is a variant employing high pass filtering to reduce thesensitivity of the receiver to 1/F noise and limits its refractoryperiod;

FIG. 7 is an alternative baseband modulation recovery structure whichcan be used in lieu of the corresponding structure of FIGS. 1 or 5; and

FIG. 8 illustrates waveforms useful in defining return-to-zero encodingformat.

DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS

FIG. 1 illustrates a demodulator which can be beneficially employed inaccordance with the present invention. FIG. 1 and other drawings hereinomit conventional apparatus such as bandpass filters (for example, abroadband filter passing 902-928 MHz), low noise amplifiers, etc. FIG. 1shows that an RF signal is fed, via antenna 10, to a pair of mixers, M1and M2. Mixers M1 and M2 are driven by a local oscillator signal (fromLO1) which is supplied in quadrature by quadrature divider 3. M1 isdriven by a 0° phase shifted LO signal on line 4, while M2 is driven by90° phase shifted LO signal on line 5. LO1 is set to a frequency that isnear to but not necessarily identical to the center frequency of thetransmitted signal, i.e. LO1 need not be phase locked or even frequencylocked to the incoming signal. The output of M1 and M2 produces bothin-phase (I-channel) and quadrature (Q-channel) baseband outputs at therespective mixers. All of the information that previously existed in thebandpass (double-sided) RF spectrum at the input to the antenna now hasbeen split into a pair of baseband signals. At the same time, phaserrotation which may exist between the transmitted signal and thefrequency of LO1 has been added to both the I and Q outputs and itsinfluence must be later removed if the desired signal (the modulatorinput signal) is to be recovered.

The outputs from each of the mixers is provided to a respective low passfilter F1 and F2. This low pass filtering in the I and Q outputs and thedirect-demodulation (homodyne) receiver takes the place of IF filteringin a conventional superheterodyne receiver. This is a potential costreduction. The I and Q baseband signals are then separately amplified(A1 and A2) and applied to respective A/D converters C1 and C2. Theoutput of the converters C1 and C2 are quantized either in single bit(i.e. by a simple comparator) process or by a multi-level process. Theresulting digitized signal is then fed into respective one chip delaysD1 and D2. The delays may be readily implemented as a multiple stageshift register whose length corresponds to a single chip to achieve a1-chip delay in the signal. The depth of the register corresponds to theresolution of the converters C1 and C2. The delayed output is thendigitally multiplied in the respective multiplier (P1 and P2) by thequantized signal to produce a pair of differentially demodulatedoutputs. Summing these outputs together in the summers produces adigital representation of the original modulator signal, with any phaserrotation that might have existed between the transmitter and receiverhaving been removed. The-output of the summer can then be de-spread by avariety of digital spread spectrum elements One suitable correlator isdescribed in the copending application Ser. No. 08/084,978, filed Jun.25, 1993. FIG. 1 shows the use of a correlator. The demodulator of FIG.1 works well with any of three different modulation formats: (1)ordinary differential encoding (DE), shown in FIG. 2; (2) differentencoding plus return-to-zero encoding (RZ/DE), shown in FIG. 3; or (3)differential encoding plus Manchester encoding (MC/DE), shown in FIG. 4.

FIG. 2 shows a modulator which can be used with the demodulator ofFIG. 1. The modulator of FIG. 2 employs differential encoding. A PNsequence generator PN cyclically generates a particular code wordcomprising n chips and provides its serial output as one input to anexclusive OR gate G1. The other input to the exclusive OR gate G1 is thedata to be transmitted. As those skilled in the art are aware, the PNsequence generator is clocked at a multiple n times the rate of the datasuch that each data bit is presented to the exclusive OR gate during theperiod when an entire code word is presented to the other input of theexclusive OR gate G1. The output of the exclusive OR gate is provided asan input to a differential encoder DE whose output is provided to amixer M3. The other input to the mixer is the output of a localoscillator LO. As is well understood in the art, the output of the mixerM3 is the output of the differential encoder modulated on the localoscillator signal. This composite signal is then transmitted via theantenna 20 and a power amplifier (not illustrated).

FIG. 3 shows another modulator which can be used with the demodulator ofFIG. 1. The modulator of FIG. 3 performs differential and RZ encoding.The only difference between the modulators of FIGS. 2 and 3 is that themodulator of FIG. 3 has a return-to-zero encoder RZE inserted betweenthe differential encoder DE and the mixer.

Because the return-to-zero format is sometimes characterizeddifferently, reference is made to FIG. 8 which shows a return-to-zeroformat. The upper line on FIG. 8 (referenced NRZ) shows a digitalwaveform in non-return-to-zero format for the bit pattern0-1-0-1-1-0-0-0-1-0-1-1-1-0. As used in this application, return-to-zerois a three-level waveform wherein a first binary signal (such as zero)is represented as a negative pulse, a second binary signal (such as aone) is represented as a positive pulse and each bit is separated fromits adjacent bit by a region of no activity or zero level. Thus the samedigital bit pattern is shown in the second line of FIG. 8 (referencedRZ) in return-to-zero format. Comparing the lines referenced NRZ and RZ,it will be appreciated that each bit time of NRZ has been divided inhalf. The first half of the bit time carries a negative or positivepulse identifying whether the particular bit is a zero or a one,respectively. The second half of each bit time, the waveform is at zerolevel. The last line of FIG. 8 (referenced DERZ) illustrates the bitpattern in differentially encoded, return-to-zero format. In thisformat, a one bit is transmitted as a pulse of the same polarity as theprevious pulse and a zero bit is transmitted as a pulse of the oppositepolarity from the previous pulse. It will be apparent from thisdefinition that the polarity of the initial pulse is arbitrary. Thoseskilled in the art will be capable of correlating these rules with thewaveforms referenced NRZ, RZ and DERZ in FIG. 8, illustrating thedigital bit stream 0-1-0-1-1-0-0-0-1-0-1-1-1-0.

FIG. 4 illustrates another modulator which can be used with thedemodulator of FIG. 1. The modulator of FIG. 4 performs differential andManchester encoding. The only difference between the modulator of FIG. 4and that of FIG. 2 is that a Manchester encoder ME is inserted betweenthe output of the differential encoder DE and the input of the mixer M3.

FIG. 5 illustrates an alternative demodulator to that of the demodulatorof FIG. 1. The only difference between the demodulator of FIG. 5 andthat of FIG. 1 is that a single A/D converter C3 is multiplexed to servethe outputs of both the I and Q channels. As shown in FIG. 5, whereasthe amplifiers A1 and A2 are associated with respective A/D convertersC1 and C2, in FIG. 1, the output of the amplifiers A1 and A2 are coupledto inputs of a single pole, double throw switch S1. The output of theswitch S1 is coupled to an A/D converter C3. The output of the A/Dconverter C3 is coupled to an input of a single poll, double throwswitch S2. The outputs of the switch S2 are inputs to the digital delaysand multipliers P1 and P2 in a manner similar to that shown in FIG. 1.The switches S1 and S2 are driven by a multiplexing signal identified asMUX. In order to perform the A/D digital conversion for both the I and Qchannels, the A/D converter C3 operates at twice the rate of A/Dconverters C1 and C2 (FIG. 1). The switches S1 and S2 operatesynchronously to share the services of the A/D converter C3, first tothe I channel (the output of the amplifier A1) and then to the Q channel(the output of the amplifier A2).

FIG. 7 shows another variation on a demodulator. Whereas the demodulatorof FIG. 1 first summed the processed I and Q channels before providingthe output to a correlator, the demodulator of FIG. 7 employscorrelators CL1 and CL2, one for the processed I channel and another forthe processed Q channel. It is the outputs of the correlators then whichare summed at S.

FIG. 6 shows a further alternative to the demodulator. FIG. 6 differsfrom the demodulator of FIG. 5 by the presence of high pass filters H1and H2, one in each of the I and Q channels. While FIG. 6 is drawn as avariant of FIG. 5, those skilled in the art will understand that thesame high pass filter structure (H1 and H2) of FIG. 6 can be employed inthe demodulator of FIG. 1 as well. The preceding discussion has pointedout that the demodulator of FIGS. 1, 5 or 7 can be paired with any ofthe modulators of FIGS. 2, 3 or 4, i.e. differential encoding,differential and RZ encoding, or differential and Manchester encoding.The demodulator of FIG. 6, however, has added a high pass filter.Inserting a high pass filter into an operable demodulator would appearto violate one of the basic precepts of proper communications systemdesign in that the bandpass shape of the receiver is no longer tailoredto match the spectrum of the transmitter, i.e. some of the transmittedenergy is discarded or not used at the receiver. It has been discovered,however, that inasmuch as the high pass filtering is a form ofdifferentiation, it is complementary to a theoretical integration at themodulator. Thus, if the modulator used Manchester coding and asubsequent integration stage, the high pass filtering in the demodulatorwould complement the integration at the modulator and result in therecovery of the original Manchester encoding. It is therefore possibleto use the demodulator of FIG. 6 (which has a "extra" high pass filterstage) in combination with a modulator operating under RZ encoding andrecover, at the demodulator, the Manchester coding corresponding to theRZ encoding actually employed at the modulator. Absent some benefitobtained by adding the high pass filtering stage, there would be nopoint in using this additional circuitry. There is, however, asignificant added benefit.

Either the demodulator of FIG. 1 or the demodulator of FIGS. 5 or 7,since they operate at what could be considered "zero IF", i.e. an IFwhich is centered at DC or the zero frequency, exhibit 1/f noise in themixers M1 and M2 as well as in the baseband amplifiers A1 and A2. Thisnoise can degrade performance. In addition, and because the basebandamplifiers couple low frequency signals, the receiver can exhibit a longsettling transient whenever it is overloaded. In time division duplexedradio systems, it is conventional to have a transmitter and receivershare a common antenna. Usually this is accomplished in an arrangementcalled a receiver/transmit or T/R switch. The T/R switch acts to diverta majority of the transmitted signal from the transmitter to the antennaand away from the receiver. However, attenuation in the T/R switchseldom exceeds 20 dB. This-means that using a 1-watt transmitter, 10milliwatts are fed directly into the receiver. For a sensitive receiver,such as those described herein, that energy can well result in atransient overload condition.

When the transient overload terminates, the receiver will have arefractory period during which it is unable to receive and properlyprocess signals. A relatively sensitive receiver must be able to detectsignals that are on the order of -90 dBm. This means that the differencebetween the transmit overload condition and the expected minimum signalis 100 dB or more. In time division multiplex and time division duplexsystems, it is desirable to have the receiver refractory period be asshort as possible. This conclusion flows from the fact that therefractory period is essentially dead time and subtracts from the datacarrying capacity of the system.

The problem of 1/f noise and refractory period recovery can be addressedby a simple high pass filtering of the baseband signals, as isillustrated in FIG. 6.

As has already been mentioned, an RZ modulated signal that is receivedand then high pass filtered will emerge from the demodulator (input tothe correlator) as a Manchester encoded signal. The Manchester code isone of the formats that is compatible with this particular demodulator.

The cutoff frequencies for the high pass filter (H1 or H2) as well asthe cutoff frequency for the low pass filter (F1, F2) can be selected independence on the data rate although those skilled in the art willunderstand that there is a range of cutoff frequencies which issuitable. Typically, the cutoff frequency for the high pass filter wouldbe approximately 50% of the transmission bit rate, whereas the cutofffrequency for the low pass filter is generally about 150% of the bitrate, for the devices described herein. For example, assuming a 500Kb/sec data rate, a suitable cutoff frequency for the high pass filterswould be about 250 KHz. A suitable cutoff frequency for the low passfilter would be about 1.25 MHz. In other applications, for example whereconventional NRZ data is transmitted, a suitable cutoff frequency forthe low pass filter would be about 50%-70% of the bit rate. For theembodiments described herein using RZ coding (which doubles thebandwidth), a higher cutoff frequency with low pass filter (e.g. the150% typical value) is desirable. As an alternative to discrete highpass and low pass filters, a suitable frequency response, in some cases,can be achieved with a single bandpass filter or filter function. Whilethis is at times suitable, it is generally more practical to place adiscrete low pass filter ahead of the baseband amplifiers and in somecases to distribute the high pass filter function through several ACcoupling points in the baseband amplifiers.

It should be apparent from a review of this application that manychanges can be made within the spirit and scope of the invention whichis to be construed in accordance with the claims appended hereto and isnot to be limited by the specific embodiments described herein.

I claim:
 1. A spread spectrum communication system including atransmitter and a receiver responsive to signals transmitted by saidtransmitter wherein:said transmitter includes means for transmittingdata in return to zero encoded form, said receiver includes means fordecoding manchester encoded data, said receiver further comprisingdifferentiation means for passing signals received from said transmitterto said means for decoding.
 2. The system of claim 1 wherein saiddifferentiation means comprises a high pass filter.
 3. The system ofclaim 2 wherein said transmitter modulates signals in spread spectrumform and said receiver includes means for despreading spread spectrummodulated signals.
 4. A communication system as recited in claim 2wherein said receiver further includes:mixer means including a localoscillator and a signal splitter responsive to said signals transmittedby said transmitter to produce baseband I and Q channel signals, saiddifferentiations means comprising a separate differentiation element foreach of said I and Q channel signals, and said means for decodingmanchester encoded data comprising a differential decoder responsive tosaid I and Q channel signals.
 5. A spread spectrum receiver for dataencoded in return-to-zero form including differentiation means forperforming a differentiation operation to produce processed signals anda manchester decoder responsive to said processed signals.
 6. A receiveras recited in claim 5 wherein said differentiation means comprises ahigh pass filter.
 7. A receiver as recited in claim 6 wherein saidencoded data is modulated in spread spectrum form and said manchesterdecoder outputs signals to despreading means for despreading signalsfrom said manchester decoder to recover said encoded data.
 8. A receiveras recited in claim 7 which further includes:mixer means including alocal oscillator and a signal splitter responsive to said signalstransmitted by said transmitter to produce baseband I and Q channelsignals, said differentiations means comprising a separatedifferentiation element for each of said I and Q channel signals, andsaid manchester decoder comprising a differential decoder responsive tosaid I and Q channel signals.
 9. A spread spectrum communication systemincluding a transmitter and a receiver responsive to signals transmittedby said transmitter wherein:said transmitter includes a manchesterencoder and an integrator for transmitting integrated, manchesterencoded data, said receiver includes means for decoding manchesterencoded data, said receiver further comprising a high pass filter forpassing signals received from said transmitter to said means fordecoding.
 10. The system of claim 9 wherein said transmitter modulatessignals in spread spectrum form and said receiver further includes meansfor despreading spread spectrum modulated signals.
 11. A communicationsystem as recited in claim 10 wherein said receiver furtherincludes:mixer means including a local oscillator and a signal splitterresponsive to said signals transmitted by said transmitter to producebaseband I and Q channel signals, said high pass filter comprising aseparate high pass filter element for each of said I and Q channelsignals, and said means for decoding manchester encoded data comprisinga differential decoder responsive to said I and Q channel signals.
 12. Amethod of communication employing direct sequence spread spectrummodulation comprising:a) modulating data for transmission using returnto zero spread spectrum modulation to produce a return to zero encodedspread spectrum modulated signal, b) transmitting the modulated signal,c) receiving the modulated signal and mixing the received signal down tobaseband, d) high pass filtering the baseband signal, e) processing thehigh pass filtered baseband signal to recover a manchester encoded formof the modulated signal, and f) despreading the manchester encoded formof the modulated signal to recover the data.
 13. A method as recited inclaim 12 which:c1) the step of receiving the modulated signal and mixingthe received signal down to baseband includes mixing the received signalwith two, phase shifted local oscillator signals to produce I and Qchannel baseband signals, d1) the step of high pass filtering thebaseband signal includes high pass filtering each of the I and Q channelbaseband signals, and e1) the step of processing the high pass filteredbaseband signal includes differential demodulation of the I and Qchannel baseband signals.